Optimized data converter design using mixed semiconductor technology for cellular communications

ABSTRACT

A cellular radio architecture for a vehicle that includes a receiver module having a receiver delta-sigma modulator that converts analog receive signals to a representative digital signal. The architecture further includes a transmitter module having a transmitter delta-sigma modulator for converting digital data bits to analog transmit signals. Portions of the receiver and transmitter modules are fabricated with indium phosphide (InP) technologies and portions of the receiver and transmitter modules are fabricated with CMOS technologies.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application claims the benefit of the priority date of U.S.Provisional Patent Application Ser. No. 62/015,272, entitled,“Optimizied Data Converter Design Using Mixed Semiconductor Technologyfor Automotive Cellular Communications”, filed Jun. 20, 2014.

BACKGROUND OF THE INVENTION

1. Field of the Invention

This invention relates generally to a cellular radio architecture for avehicle and, more particularly, to a cellular radio architecture for avehicle that includes a receiver module and a transmitter module, whereportions of the receiver and transmitter modules are fabricated withindium phosphide (InP) technologies and portions are fabricated withCMOS technologies.

2. Discussion of the Related Art

Traditional cellular telephones employ different modes and bands ofoperations that have been supported in hardware by having multipledisparate radio front-end and baseband processing chips integrated intoone platform, such as tri-band or quad-band user handsets supportingGSM, GPRS, etc. Known cellular receivers have integrated some of theantenna and baseband data paths, but nevertheless the current state ofthe art for mass mobile and vehicular radio deployment remains amultiple static channelizing approach. Such a static architecture iscritically dependent on narrow-band filters, duplexers andstandard-specific down-conversion to intermediate-frequency (IF) stages.The main disadvantage of this static, channelized approach is itsinflexibility with regards to the changing standards and modes ofoperation. As the cellular communications industry has evolved from 2Gto 3G, 4G and beyond, each new waveform and mode has required a redesignof the RF front-end of the receiver as well as expanding the basebandchip set capability, thus necessitating a new handset. For automotiveapplications, this inflexibility to support emerging uses isprohibitively expensive and a nuisance to the end-user.

Providing reliable automotive wireless access is challenging from anautomobile manufacturers point of view because cellular connectivitymethods and architectures vary across the globe. Further, the standardsand technologies are ever changing and typically have an evolution cyclethat is several times faster than the average service life of a vehicle.More particularly, current RF front-end architectures for vehicle radiosare designed for specific RF frequency bands. Dedicated hardware tunedat the proper frequency needs to be installed on the radio platform forthe particular frequency band that the radio is intended to operate at.Thus, if cellular providers change their particular frequency band, theparticular vehicle that the previous band was tuned for, which may havea life of 15 to 20 years, may not operate efficiently at the new band.Thus, this requires automobile manufactures to maintain a myriad ofradio platforms, components and suppliers to support each deployedstandard, and to provide a path to upgradability as the cellularlandscape changes, which is an expensive and complex proposition.

Known software-defined radio architectures have typically focused onseamless baseband operations to support multiple waveforms and haveassumed similar down-conversion-to-baseband specifications. Similarly,for the transmitter side, parallel power amplifier chains for differentfrequency bands have typically been used for supporting differentwaveform standards. Thus, crucially, receiver front-end architectureshave typically been straight forward direct sampling or one-stage mixingmethods with modest performance specifications. In particular, no priorapplication has required a greater than 110 dB dynamic range withassociated IP3 factor and power handling requirements precisely becausesuch performance needs have not been realizable with complementary metaloxide semiconductor (CMOS) analog technology. It has not been obvioushow to achieve these metrics using existing architectures for CMOSdevices, thus the dynamic range, sensitivity and multi-mode interleavingfor both the multi-bit analog-to-digital converter (ADC) and thedigital-to-analog converter (DAC) is a substantially more difficultproblem.

Software-defined radio architectures do not exist in the automotivedomain, but have been proposed and pursued in other non-automotiveapplications, such as military radios with multi-band waveforms.However, in those arenas, because of vastly different waveform needs,conflicting operational security needs and complex interoperabilityrequirements, a zero-IF approach has proven technically difficult. Knownsoftware defined radios have typically focused on backend processing,specifically providing seamless baseband operations to support multiplewaveforms. The modest performance specifications haven't demandedanything more aggressive from front-end architectures. Straight-forwarddirect sampling or 1-stage mixing methods have been sufficient in thereceiver. For software defined radios that employ delta-sigmamodulators, the component function is commonly found after a downconversion stage and has low-pass characteristics. With regard to thetransmitter, parallel multiple power amplifier chains to supportdiffering frequency bands and waveform standards have been sufficientfor meeting the requirements.

SUMMARY OF THE INVENTION

The present disclosure describes a cellular radio architecture for avehicle that includes a triplexer coupled to an antenna structure andincluding three signal paths, where each signal path includes a bandpassfilter that passes a different frequency band than the other bandpassfilters and a circulator that provides signal isolation between thetransmit signals and the receive signals. The architecture also includesa receiver module having a separate signal channel for each of thesignal paths in the triplexer, where each signal channel in the receivermodule includes a receiver delta-sigma modulator that converts analogreceive signals to a representative digital signal, and where portionsof the receiver module are fabricated with indium phosphide (InP)technologies and portions of the receiver module are fabricated withCMOS technologies. The architecture further includes a transmittermodule having a transmitter delta-sigma modulator for converting digitaldata bits to analog transmit signals, where the transmitter moduleincludes a power amplifier and a switch for directing the analogtransmit signals to one of the signal paths in the triplexer, and whereportions of the transmitter module are fabricated with indium phosphide(InP) technologies and portions of the transmitter module are fabricatedwith CMOS technologies.

Additional features of the present invention will become apparent fromthe following description and appended claims, taken in conjunction withthe accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram of a known multi-mode, multi-band cellularcommunications handset architecture;

FIG. 2 is a block diagram of a software-programmable cellular radioarchitecture applicable for automotive uses;

FIG. 3 is a block diagram of a receiver channel for the radioarchitecture shown in FIG. 2 including a delta-sigma modulator showinginterleaver quantizers and a transmitter module;

FIG. 4 is a schematic diagram of a 4-bit quantizer employed in thedelta-sigma modulator shown in FIG. 3;

FIG. 5 is a schematic block diagram of the sixth-order filter employedin the delta-sigma modulator shown in FIG. 3;

FIG. 6 is a schematic diagram of a tuneable transconductance amplifieremployed in the sixth-order filter shown in FIG. 5;

FIG. 7 is a capacitor array providing course and fine tuning employed inone of the resonators in the sixth-order filter shown in FIG. 5;

FIG. 8 is a schematic block diagram of a digital bandpass delta-sigmamodulator for the transmitter module shown in FIG. 3;

FIGS. 9-11 show three embodiments of a delta-sigma modulator employingan interleaving DEM algorithm for the transmitter module of the radioarchitecture shown in FIG. 3;

FIG. 12 is a schematic diagram of a timing circuit that can be employedin the delta-sigma modulator shown in FIG. 3; and

FIG. 13 is a profile view of a semiconductor device showing integrationbetween CMOS and InP technologies.

DETAILED DESCRIPTION OF THE EMBODIMENTS

The following discussion of the embodiments of the invention directed toa cellular radio architecture for a vehicle is merely exemplary innature, and is in no way intended to limit the invention or itsapplications or uses. For example, as mentioned, the radio architectureof the invention is specific for a vehicle. However, as will beappreciated by those skilled in the art, the radio architecture may haveother applications other than automotive applications.

FIG. 1 is a block diagram of a known multi-mode, multi-band cellularcommunications user handset architecture 10 for a typical cellulartelephone. The architecture 10 includes an antenna structure 12 thatreceives and transmits RF signals at the frequency band of interest. Thearchitecture 10 also includes a switch 14 at the very front-end of thearchitecture 10 that selects which particular channel the transmitted orreceived signal is currently for and directs the signal through adedicated set of filters and duplexers represented by box 16 for theparticular channel. Modules 18 provide multi-mode and multi-band analogmodulation and demodulation of the receive and transmit signals andseparates the signals into in-phase and quadrature-phase signals sent toor received from a transceiver 20. The transceiver 20 also convertsanalog receive signals to digital signals and digital transmit signalsto analog signals. A baseband digital signal processor 22 provides thedigital processing for the transmit or receive signals for theparticular application.

FIG. 2 is a schematic block diagram of a cellular radio front-endarchitecture 30 that provides software programmable capabilities as willbe discussed in detail below. The architecture 30 includes an antennastructure 32 capable of receiving and transmitting the cellulartelephone frequency signals discussed herein, such as in a range of 400MHz-3.6 GHz. Signals received and transmitted by the antenna structure32 go through a triplexer 34 that includes three signal paths, whereeach path is designed for a particular frequency band as determined by abandpass filter 36 in each path. In this embodiment, three signal pathshave been selected, however, the architecture 30 could be expanded toother numbers of signal paths. Each signal path includes a circulator 38that separates and directs the receive and transmit signals and providesisolation so that the high power signals being transmitted do not enterthe receiver side and saturate the receive signals at those frequencybands.

The architecture 30 also includes a front-end transceiver module 44 thatis behind the triplexer 34 and includes a receiver module 46 thatprocesses the receive signals and a transmitter module 48 that processesthe transmit signals. The receiver module 46 includes three receiverchannels 50, one for each of the signal paths through the triplexer 34,where a different one of the receiver channels 50 is connected to adifferent one of the circulators 38, as shown. Each of the receiverchannels 50 includes a delta-sigma modulator 52 that receives the analogsignal at the particular frequency band and generates a representativestream of digital data using an interleaving process in connection witha number of 4-bit quantizer circuits operating at a very high clockrate, as will be discussed in detail below. As will further bediscussed, the delta-sigma modulator 52 compares the difference betweenthe receive signal and a feedback signal to generate an error signalthat is representative of the digital data being received. The digitaldata bits are provided to a digital signal processor (DSP) 54 thatextracts the digital data to provide the audio content in the receivesignal. A digital baseband processor (DBP) 56 receives and operates onthe digital data stream for further signal processing in a manner wellunderstood by those skilled in the art. The transmitter module 48receives digital data to be transmitted from the processor 56. Themodule 48 includes a transmitter circuit 62 having a delta-sigmamodulator that converts the digital data to an analog signal to beamplified by a power amplifier 64, as will be discussed in detail below.The amplified analog signal from the power amplifier 64 is then sent toa switch 66 that directs the signal to the particular circulator 38 inthe triplexer 34 depending on which frequency is being transmitted.

As will become apparent from the discussion below, the configuration ofthe architecture 30 provides software programmable capabilities throughhigh performance delta-sigma modulators that provide optimizedperformance in the signal band of interest and that can be tuned acrossa broad range of carrier frequencies. The architecture 30 meets currentcellular wireless access protocols across the 0.4-2.6 GHz frequencyrange by dividing the frequency range into three non-continuous bands.However, it is noted that other combinations of signal paths andbandwidth are of course possible. The triplexer 34 implements frequencydomain de-multiplexing by passing the RF carrier received at the antennastructure 32 into one of the three signal paths. Conversely, thetransmit signal is multiplexed through the triplexer 34 onto the antennastructure 32. For vehicular wireless access applications, such alow-cost integrated device is desirable to reduce parts cost,complexity, obsolescence and enable seamless deployment across theglobe.

The circulators 38 route the transmit signals from the transmittermodule 48 to the antenna structure 32 and also provides isolationbetween the high power transmit signals and the receiver module 46.Although the circulators 38 provide significant isolation, there is someport-to-port leakage within the circulator 38 that provides a signalpath between the transmitter module 48 and the receiver module 46. Asecond undesired signal path occurs due to reflections from the antennastructure 32. As a result, a portion of the transmit signal will bereflected from the antenna structure 32 due to a mismatch between thetransmission line impedance and the antenna's input impedance. Thisreflected energy follows along the same signal path as the incomingdesired signal back to the receiver module 46. The impact of thisimperfect isolation and antenna reflection has traditionally beenresolved through the use of static surface acoustic wave (SAW) or bulkacoustic wave (BAW) filters. However, these types of filters aregenerally employed for fixed frequencies and do not support areconfigurable radio architecture.

The delta-sigma modulators 52 are positioned near the antenna structure32 so as to directly convert the RF receive signals to bits in thereceiver module 46 and bits to an RF signal in the transmitter module48. The main benefit of using the delta-sigma modulators 52 in thereceiver channels 50 is to allow a variable signal capture bandwidth.This is possible because the architecture 30 enables softwaremanipulation of the modulator filter coefficients to vary the signalbandwidth and tune the filter characteristics across the RF band, aswill be discussed below.

The architecture 30 allows the ability to vary signal capture bandwidth,which can be exploited to enable the reception of continuous carrieraggregated waveforms without the need for additional hardware. Carrieraggregation is a technique by which the data bandwidths associated withmultiple carriers for normally independent channels are combined for asingle user to provide much greater data rates than a single carrier.Together with MIMO, this feature is a requirement in modern 4G standardsand is enabled by the OFDM family of waveforms that allow efficientspectral usage.

The architecture 30 through the delta-sigma modulators 52 can handle thesituation for precise carrier aggregation scenarios and bandcombinations through software tuning of the bandpass bandwidth, and thusenables a multi-segment capture capability. Dynamic range decreases forwider bandwidths where more noise is admitted into the samplingbandpass. However, it is assumed that the carrier aggregation typicallymakes sense when the user has a good signal-to-noise ratio, and not cellboundary edges when connectivity itself may be marginal. Note that theinter-band carrier aggregation is automatically handled by thearchitecture 30 since the triplexer 34 feeds three independentmodulators in the channels 50.

The architecture 30 is also flexible to accommodate other wirelesscommunications protocols. For example, a pair of switches 40 and 42 canbe provided that are controlled by the DSP 56 to direct the receive andtransmit signals through dedicated fixed RF devices 58, such as a GSM RFfront end module or WiFi FEM. In this embodiment, some select signalpaths are implemented via conventional RF devices. FIG. 2 only shows oneadditional signal path, however, this concept can be expanded to anynumber of additional signal paths depending on use cases and services.

FIG. 3 is a schematic block diagram 68 of a portion of the architecture30 including one of the receiver channels 50 having the delta-sigmamodulator 52 and the DSP 54, the transmitter module 48 and the basebandprocessor 56. The receive signals from the circulator 38 at node 92 areprovided to a broadband combiner 70 operating as a summation junction. Areceive feedback signal on line 94 and a transmitter cancellation signalon line 96 are also provided to the combiner 70 and are subtracted fromthe receive signal to generate an analog error signal that is sent to alow noise transconductance amplifier (LNTA) 72. The amplified errorsignal is provided to a sixth-order LC filter 74, where the filter 74operates as a bandpass filter to reshape the noise in the analog errorsignal so that it is out of the desired signal band, thus obtainingquality signal-to-noise and distortion performance.

In low-pass or low-IF bandwidth modular designs having a faster clockrate means a higher oversampling ratio (OSR), i.e., the ratio of theclock speed to twice the signal bandwidth, which means better dynamicrange. In a high-RF bandpass design, the clock rate is constrained bythe clock-to-carrier ratio. For a sampling rate to RF carrier frequencyratio less than four, the digital filter becomes substantially morecomplicated. To meet cellular standards in the high receive band up to2.6 GHz, a clock rate of 10.4 GHZ would be required. However, for theseclock rates, achieving 100 dB of dynamic range in the analog feedbacksignal to the combiner 70 is unrealistic. To address this challenge, thepresent invention provides interleaved quantizer circuits to reduce theclock rate to 5.2 GHz. The quantizer circuits are clocked at half rate,but the achieved clock rate is still 10.4 GHz, which has the advantageof maintaining a large OSR greater than 512 for a 20 MHz signal band,but making stability more challenging since the feedback delay isrelative to the effective clock period it is easier to exceed athreshold for stability.

Based on this discussion, the filtered error signal from the filter 74is provided to a series of multi-bit ADCs 76, such as two or more ADCswith three or four-bit resolution, that convert the error signal to adigital signal in an interleaving manner at the output of the ADC 76.Interleaving as used herein means that the analog carrier frequency fromthe filter 74 is processed in separate sections by the plurality of theADCs 76 so that the rate at which the conversion process is beingperformed can be reduced. Those bits are sent to a series of multi-bitDACs 78 having, for example, three or four-bit resolution in thefeedback line 94, where the combination of each pair of the ADC 76 andthe DAC 78 is a quantizer circuit that operates as a 4-bit interleaveron multiple groups of 4-bits, as will be discussed in further detailbelow. A bit resolution of four was chosen to balance the challenge indesign of the LNTA 72 and the feedback DACs 78. An upper bound on thenumber of bits is set by the ability to design the DAC 78 to meet thedynamic range of the system. Unlike the 4-bit ADC that is in the forwardpath of the modulator 52, the non-linearities of the DAC 78 are notshaped by the filter 74 and directly affect the performance. Theinterleaving process provides the groups of 4-bits from the ADCs 76through a data weighted averaging (DWA) digital shaper circuit 86 beforebeing provided to the 4-bit DACs 78. Because interleaving allows thesampling rate to be reduced oversampling to maintain bit integrity isnot required. Thus the bit resolution is four, but each pair of the ADCs76 and the DACs 78 in the quantizer circuit need only operate athalf-speed making it easier to meet dynamic range requirements.

Interleaving both the multi-bit ADC 76 and DAC 78 as proposed herein isa substantially harder problem than addressed in the prior art. Meetingdynamic range goals requires matching among the interleaved ADCs 76 andthe DACs 78 in addition to managing mismatch within an individual DAC.Also, interleaving increases the excess phase delay in the loop andrequires compensation to maintain stability. The clock rates indelta-sigma ADC designs used herein are typically as fast as possiblewithout degrading the modulator performance by introducing jitter intothe system.

FIG. 4 is a schematic diagram of a quantizer circuit 100 defined by onegroup of the ADCs 76 and the DACs 78. The ADC 76 and the DAC 78 includea bit path for each bit in the 4-bit quantizer. Each signal path in theADC 76 includes a comparator 102 and a latch 104. A voltage dividernetwork 106 sets a different reference voltage for each of thecomparators 102 in each of the signal paths. The analog signal from thefilter 74 is provided on line 108 and is sent to each of the comparators102. If that voltage level is above the reference level for theparticular comparator 102, then the latch 104 for that channel is sethigh for that bit. The DAC 78 includes latches 110 that reset the bitsback to an analog signal through a switch 112.

Delta-sigma modulators are a well known class of devices forimplementing analog-to-digital conversion. The fundamental propertiesthat are exploited are oversampling and error feedback (delta) that isaccumulated (sigma) to convert the desired signal into a pulse modulatedstream that can subsequently be filtered to read off the digital values,while effectively reducing the noise via shaping. The key limitation ofknown delta-sigma modulators is the quantization noise in the pulseconversion process. Delta-sigma converters require large oversamplingratios in order to produce a sufficient number of bit-stream pulses fora given input. In direct-conversion schemes, the sampling ratio isgreater than four times the RF carrier frequency to simplify digitalfiltering. Thus, required multi-GHz sampling rates have limited the useof delta-sigma modulators in higher frequency applications. Another wayto reduce noise has been to use higher order delta-sigma modulators.However, while first order canonical delta-sigma architectures arestable, higher orders can be unstable, especially given the tolerancesat higher frequencies. For these reasons, state of the art higher orderdelta-sigma modulators have been limited to audio frequency ranges,i.e., time interleaved delta-sigma modulators, for use in audioapplications or specialized interleaving at high frequencies. Thepresent invention improves upon prior approaches through the sixth-orderfilter 74 with the 4-bit feedback structure for maximum flexibility inthe noise shaping characteristics. The modulators 52 can achieve adynamic range of 100 dB over a signal bandwidth of 20 MHz across an RFbandwidth of 400 MHz.

A decoder 82 receives the 4-bit sequence from all of the ADCs 76simultaneously and reconfigures the bits in the proper orientation to beoutput as a serial data stream. The bits are then provided to ade-multiplexer 84 to provide the data stream at the lower clock rate.Particularly, the output data bits from the ADCs 76 are decoded fromthermometer code to binary code and de-multiplex down to a data ratethat can be supported by the DSP 54. Operational parameters for thefilter 74 are set by the DSP 54 and are provided on line 98. Also, acalibration phase is performed to optimize the performance for thereceive channel 50. A clocking device 88 provides the clock signals tothe various components in the modulator 52, as shown, where thefrequency of the clock rate determines power consumption andsemiconductor material as will be discussed in further detail below.

FIG. 5 is a schematic diagram of the sixth-order filter 74 that includesthree passive LC resonator circuits 120 each including an inductor 122and a capacitor array 114. As is well understood by those skilled in theart, the number of orders of a particular filter identifies the numberof poles, where the number of poles defines the number of LC circuits.Filtering and correction algorithms are applied to the digital outputstream in the DSP 54. Low-speed tuning and calibration signals are fedback into the filter 74 on the line 98 for reconfiguring and optimizingthe filter 74. The filter 74 also includes a series of transconductanceamplifiers 126 in the primary signal path. A feed-forward path providesstability and includes integrator circuits 128 each including anintegrator 130, two tuneable transconductance amplifiers 132 and 134 anda summer 136. The filter 74 has a feed-forward architecture wheresignals are fed from early resonator stages into a final summingjunction 138. The signal from the LTNA 72 is provided at node 140, andthe most significant bit of the 4-bit DAC 78 in the feedback loop on theline 94 is provided at node 142. The output of the filter 74 is providedto the 4-bit ADCs 76 at node 144. A low-speed DAC array 146 receivescoefficient control bits at node 148 from the line 98 to control thefunctions in the integrator circuits 128. Frequency control bits fromthe DSP 54 on the line 98 are provided to the capacitor array 124 atnode 150.

Although a feedback architecture may offer more out-of-band noiseshaping, a feed-forward architecture gives more flexibility in designinga stable modulator. The passive resonator circuits 120 are employedbecause they have lower noise figures, higher linearity, require lesspower, and can operate at higher carrier frequencies than activeresonators. The quality factor Q of the resonator circuits 120 isprimarily set by the series resistance in the inductor 122. Simulationshave shown that the resonator Q should be greater than 30 to achieve adeep notch characteristic and will require an off-chip component as itis difficult to achieve the needed performance with an integrated spiralinductor. Simulations also have shown that five sets of coefficients areneeded to cover a 400 MHz RF band and the modulator 52 is stable across100 MHz band, but an extra set allows for frequency overlap. Thecoefficient set information will be stored in a look-up table in the DSP54. Control bits will be loaded and sent to the filter 74 to set thenotch frequency and component parameters based on RF carrierinformation.

FIG. 6 is a schematic diagram of an amplifier circuit 160 implemented inthe transconductance amplifiers 126 to show how the coefficients may beadjusted. The gain of the transconductance amplifiers 126 may be variedby applying a differential voltage to a cross-coupled pair oftransistors 162. When the differential voltage is zero, there is nogain. For a large positive differential voltage, the amplifier 126achieves a maximum positive gain and, conversely, for large negativedifferential voltage, the transconductance gain is maximized andinverted. The differential voltages are converted from the DSP controlbits using the low-speed DAC array 146.

The inductors 122 will be fixed for each of the three RF bands in thesignal paths through the triplexer 34 and the capacitance changed withineach band through the capacitor array 124. The size, or moreappropriately, the resolution of the capacitor array 124 will be fineenough for narrow frequency selectivity. FIG. 7 is a schematic diagramof one possible embodiment for the capacitor array 124 that providescoarse and fine tuning. The capacitor array 124 includes a plurality ofcapacitors 170 and switches 172, where the switches 172 are selectivelycontrolled by the frequency notch control signal from the DSP 54 at node176.

The order of the filter 74, the ratio of the sample rate to signalbandwidth and the number of bits in the quantizer circuit 100 are allchosen to achieve a 100 dB dynamic range. The passive resonator circuits120 with the inductors 122 and the capacitor arrays 124 offerlower-power and wider bandwidth operation. Post-processing, such asfiltering, calibration and correction of the output digital data isperformed in the DSP 54. The modulators 52 are able to meet the dynamicrange requirements by reducing the magnitude of the unwanted transmitsignal.

Additional corrections are provided on the feedback line 96 for reducinga potential interfering signal from the transmitter module 48 includinga 4-bit delta-sigma DAC 80 to replicate the transmit signal from adigital data stream and subtracted from the input in the combiner 70. Ifa transmit signal is occurring in the same frequency range through thesame circulator 38 while the delta-sigma modulator 52 is receiving areceive signal on that channel, the transmit signal is also fed back tothe combiner 70 through the delta-sigma DAC 80. The bit sequence for thetransmit signals provided by the transmitter module 48 is sent through alatch 90 that latches the bits into the DAC 80.

The adaptive cancellation technique leverages the fact that the digitalsequence for creating the transmit signal is available and uses theinherent feedback path of the modulator 52. The transmit digital datafrom the baseband processor 56 is a modified sequence of the actualtransmitted signal and has been altered based on the changingenvironment so that the replicated transmit signal, particularly thephase and strength, is a better approximation to the unwanted signalarriving at the receiver input. The modified transmit data sequence isthen converted to an analog signal through the 4-bit replica DAC 80. Theamount of cancellation needed is determined by the linearityspecification of the LNTA 72.

For a transmit power level of 25 dBm the reflected signal from theantenna structure 32 into the receiver module 46 will be approximately14 dBm, where a 3:1 VSWR and 1 dB cable loss are assumed. In traditionaldelta-sigma modulator designs, the combiner 70 comes after the LNTA 72.In a traditional architecture where the LNTA 72 is in front of thecombiner 70, the input intercept point of the LNTA 72 would have to be64 dBm to reach 100 dB of dynamic range. Since the LNTA 72 must alsoachieve a noise figure of less than 2.5 dB, this design is not feasible.The dynamic range of the receiver module 46 is heavily challenged byhaving to detect a very weak desired signal in the presence of theradios large transmitted signal. Less than ideal antenna reflections andimperfect transmit-to-receive isolation may present a fairly largetransmitted signal at a frequency near the smaller desired signalfrequencies.

If the combiner 70 precedes the LNTA 72, then the signal presented tothe LNTA 72 is the delta between the input signal and its quantizedestimate. Reducing the signal level into the LNTA 72 eases the linearityrequirements on the system for potentially higher dynamic range in theoverall modulator. The error is reduced by 6 dB for every bit in thefeedback DAC 78. The present invention proposes features for enhancinglinearity in the delta-sigma modulator 52 including having the combiner70 precede the LNTA 72 and using an adaptive cancellation scheme toattenuate the interfering transmitted signal as discussed. Swapping theorder of the combiner 70 and the LNTA 72 and using the 4-bit feedbackDACs 78, the input into the LNTA 72 is reduced to −10 dBm. The requiredamount of cancellation from the adaptive scheme needs to be at least 15dB to reduce the design requirements on the LNTA 72 to an achievable P3of 25 dBm.

For the cellular application discussed herein that covers multipleassigned frequency bands, a transmitter with multi-mode and multi-bandcoverage is required. Also, many current applications mandatetransmitters that rapidly switch between frequency bands during theoperation of a single communication link, which imposes significantchallenges to typical local oscillator (LO) based transmitter solutions.This is because the switching time of the LO-based transmitter is oftendetermined by the LO channel switching time under the control of theloop bandwidth of the frequency synthesizer, around 1 MHz. Hence, theachievable channel switching time is around several microseconds, whichunfortunately is too long for an agile radio. A fully digital PWM basedmulti-standard transmitter, known in the art, suffers from highdistortion, and the channel switching time is still determined by the LOat the carrier frequency. A DDS can be used as the LO sourced to enhancethe switching speed, however, this design consumes significant power andmay not deliver a high frequency LO with low spurious components.Alternately, single sideband mixers can be used to generate a number ofLOs with different center frequencies using a common phase-lock loop(PLL), whose channel switching times can be fast. However, this approachcan only support a limited number of LO options and any additionalchannels to cover the wide range of the anticipated 4G bands would needextra mixtures. As discussed, sigma-delta modulators have been proposedin the art to serve as an RF transmitter to overcome these issues.However, in the basic architecture, a sigma-delta modulator cannotprovide a very high dynamic range in a wideband of operations due to amoderate clock frequency. It is precisely because the clock frequency isconstrained by current technology that this high frequency mode ofoperations cannot be supported.

Returning to FIG. 3, the transmitter module 48 includes a multiplexer180 that receives the digital data to be transmitted from the basebandprocessor 56 and a delta-sigma modulator 182 that modulates the bits inthe manner as discussed herein. The modulated bits are then provided toa DWA circuit 184 and the bits are converted to an analog signal by a4-bit DAC 188. The analog signal is then amplified by the poweramplifier 64 and switched into the appropriate signal path in thetriplexer 34 by the switch 66. A DSP 190 receives a signal from the DWAcircuit 184 to provide a representation of the transmit signal to theDAC 80 for signal cancellation in the combiner 70 as discussed above.The multiplexer 180, the delta-sigma modulator 182, the DWA circuit 184,the DAC 188 and the DSP 190 are all part of the transceiver circuit 62.

FIG. 8 is a schematic block diagram 200 of a portion of the transmittermodule 48 showing the delta-sigma modulator 182, the DWA circuit 184 andthe DAC 188. The DWA circuit 184 modulates the digital thermal codes toshape out voltage and timing mismatches among DAC weighting elements 202that are controlled by a shape controller 204. The weighted digital bitsare then provided to the 4-bit DAC 188 that generates the analog signalto be transmitted.

The delta-sigma modulator 182 employed as an RF transmitter providesdigital data that can be generated by a high speed processor or can beproduced by a multi-rate digital signal processor. The interleavingarchitecture effectively increases the clock rate of the delta-sigmamodulator, boosts the oversampling ratio, and in turn improves theachievable signal-to-noise ratio and dynamic range. In order to enablethis interleaving architecture, an interleaving dynamic element matching(DEM) algorithm must be employed. Unlike conventional DEM algorithmsthat arrange the cells in one DAC, the interleaving DEM algorithmconsiders the used cells in all of the interleaving DACs, and arrangesthem to ensure there is no periodic pattern when using the cell.

FIG. 9 is a schematic block diagram 230 of a portion of the transmittermodule 48 showing the delta-sigma modulator 182, the DAC 188, and thepower amplifier 64 illustrating an interleaving architecture. The DWAcircuit 184 is not shown for clarity purposes. The digital signal to betransmitted from the baseband processor 56 is provided to a number ofDEM circuits 232 in the delta-sigma modulator 182 to provide the dynamicelement matching control provided by an interleaving control processor234 at the slower clock rate. The signals are combined by a summer 236and sent to the DAC 188 and then to the power amplifier 64. The DEMalgorithm operating in the circuits 232 does not run at the highestclock rate, but instead the computation is distributed into the multiplecircuits 232 running at a slower clock rate. The digital data is thenmultiplexed into one high speed data stream and fed into high speed datafollowed by the power amplifier 64.

FIG. 10 is a schematic block diagram 240 of a portion of the transmittermodule 48 that can replace the block diagram 230, where like elementsare identified by the same reference number. In this design, threeseparate DACs 242, one for each of the DEM circuits 232, replace the DAC188, where the summation junction 236 is provided after the DACs 242.

FIG. 11 is a schematic block diagram 250 of a portion of the transmittermodule 48 that can replace the block diagram 240, where like elementsare identified by the same reference number. In this design, threeseparate power amplifiers 292, one for each of the DEM circuits 232,replace the power amplifier 64.

The architecture 30 is designed to achieve 100 dB of dynamic range over20 MHz signal band and RF carrier frequencies up to 2.6 GHz. Themodulators 52 must be capable of detecting a small desired signal, suchas −86 dBm, in the presence of a large unwanted interference signal,such as 14 dBm. However, when the operating scenario does not demand asmuch from the hardware, i.e., smaller signal bandwidths, the powerdissipation in the transmitter module 48 should be reduced. There aretwo primary approaches for adjusting power dissipation includingreducing functionality or reducing performance. Reducing functionalityis a matter of reconfiguring the architecture to bypass or disableunneeded circuitry. Reducing performance includes modifying thearchitecture 30 to operate with decreased supply current or voltagethereby trading off performance for power. As will be discussed below,the present invention proposes several techniques in both of thesecategories that may be used to reduce power in the software-programmablecellular radio as discussed herein.

A first power reduction technique includes reducing the order of thefilter 74 in situations where full dynamic range is not required.Particularly, the sixth-order LC filter 74 can be reduced to afourth-order or a second-order filter by powering down followingresonator stages. For example, by disabling the last group of theresonator circuit 120, the amplifier 126 and the integrator circuit 128,represented by dotted box 228 in FIG. 5, would reduce the filter 74 froma sixth-order filter to a fourth-order filter. A separate power supplymay be used to completely shut down some circuitry while other circuitrycould remain on, but at minimal power consumption.

A second power reduction technique is to reduce the bit resolution ofthe quantizer circuit 100 from 4-bits to one bit. For this powerreduction technique, only one of the bits in the 4-bit quantizer circuit100 is employed to reduce power. For example, a center one of thecomparators 102 in the circuit 100 determines the zero cross-over pointand is required for the one-bit operation. All of the other comparators102 and most of the latches 104 and 110 may be turned off. All of theDAC current switches must remain active and be driven by the mostsignificant bit (MSB) path to keep the modulator stable. For either theone-bit or the 4-bit operation, the tail currents in the switches may bedecreased when the receive signal is not large, such as when the radiois not transmitting.

Varying the clock rate to reduce static power dissipation is anotherproposed technique to reduce power consumption. The impact on thedelta-sigma ADC architecture for this technique includes that at aslower clock rate the quantizer circuits do not need to be interleaved,and fewer stages of de-multiplexing is required. The only reasoninterleaving is provided is to support fast sampling rates for high RFcarrier frequencies. At lower RF carrier frequencies, the clock rate maybe reduced and the interleaved quantizers disabled. If the clock rate issufficiently slow, a one-to-two (1×2) demultiplexer may be all that isneeded to interface with the DSP 54 as will be described below.

FIG. 12 is a schematic block diagram of a 1×4 demultiplexer circuit 260that can be reconfigured as a 1×2 demultiplexer circuit to reduce powerdissipation for slower clock rate operation the clock rate in the mannerdiscussed herein. The demultiplexer circuit 260 receives the data to beclocked in at node 262 and a one-half divided clock signal at node 264.The data is provided to a 1×2 demultiplexer 266 and a selector 268,where the demultiplexer 266 is clocked at the one-half clock signal. Theselector 268 selects the normal data at the node 264 or thedemultiplexed data from the demultiplexer 266, and outputs the selecteddata to a 1×2 demultiplexer 270. The one-half clock signal at the node264 is divided by two by a divider 272 to generate a one-fourth clocksignal that clocks the demultiplexer 270 and a demultiplexer 274.Outputs from the demultiplexers 270 and 274 are provided to a DSP 276.When the normal data is selected, the demultiplexers 266 and 274 neednot be powered.

In other power-saving techniques, the present invention proposes tradingperformance for power consumption in relaxed operation scenarios. As anexample, in a maximum performance mode the intercept point of the LNTA72 has about a 25 dBm to support 100 dB of dynamic range while achievinga low-noise figure. If the constraint on linearity or noise can beeased, then a substantial amount of power can be saved. Two scenariosare considered to accomplish this. In the first scenario, the receivesignal is well above the noise floor and the required dynamic range isless. In the second scenario, there is no unwanted transmitted signal sothe maximum expected input level will be lower. In both scenarios thereis a relief in design for a high third-order intercept point that can betranslated to reduced current, supply voltage or both.

Performance may also be traded for power savings in the DACs 78. Sinceerrors in the DAC 78 are not shaped by the filter 74, the DACperformance must equal or exceed the modulator performance. To achievethis performance, dynamic element matching provided by the DEM circuits232 is incorporated in the DAC design as discussed above. Mismatchesamong nominally identical circuit elements inevitably introduced duringcircuit fabrication cause non-linear distortion. By scrambling the usagepattern of the elements, the DEM circuits 232 cause the error resultingfrom the mismatches to be pseudo-random noise that is uncorrelated withthe input sequence instead of non-linear distortion. If operationconditions require less dynamic range, a lower-power simplifiedscrambler would be sufficient.

Another proposed method for programmable power efficiency in the radioarchitecture 30 includes disabling the transmit cancellation scheme. Thecancellation scheme is implemented in part by the 4-bit DAC 80 forreducing self-interference. Cancellation is only necessary if thetransmit signal is in an adjacent band, is at full output power, and thereflection from the antenna structure 32 is poor. Under theseconditions, there must be cancellation so that the modulator 52 canlinearly process this unwanted interference as it appears at thereceiver module 46 so that the DSP function can process it further. Thedelta-sigma modulator 52 may be programmed to employ any combinations ofthe techniques for optimizing power efficiency in the transmitter of anautomotive wireless cellular communications system. An importantscenario is when the transceiver module 44 is in an idle state and allof the power-saving techniques are in effect. In such a scenario, thedelta-sigma modulator 52 will require only minimal functionality.

Although the RF industry has rapidly progressed with regard to compactradio architectures, existing front-end components, such as poweramplifiers, low noise amplifiers and filters still limit the bandwidthin dynamic range of these components. A single RF front-end capable ofwide bandwidth sampling has been contemplated before, but the devicetechnology was not sufficiently developed to allow the design andintegration of a multi-function radio that would be suitable as acellular handset. As discussed, an integrated front-end RF module wouldneed 111-125 dB of dynamic range for 20 MHz of signal bandwidth. CMOScannot come close to this requirement and is moving in the wrongdirection. SiGe technology is still far away, and GaAs technology isgetting closer, but still falls short.

To overcome these limitations, the present invention leverages threeunique innovations as discussed above, namely, an inherently widebandarchitecture with direct sampling using delta-sigma modulators, highlylinear based power amplifiers and input transconductor amplifiers, andtunable/programmable filters. Based on these innovations, some of thecomponents of the architecture 30 will be fabricated in indium phosphide(InP) technologies to provide the desired performance and power handlingand some of the components will be fabricated in the CMOS technology,which is lower cost. The present invention proposes that the low-powerdelta-sigma modulators 52 incorporate an InP DHBT design and fabricationprocesses and provide 200 GHz FMAX that provides sufficient head room toenable new feedback linearization techniques. Further, on thetransmitter side, InP DHBT provides a system design flexibility overother technologies. The proposed software defined front-end transceivermodule 44 is enabled by tightly integrating InP technology with silicon(Si) CMOS. Generally, those devices, components and devices that operateat the higher frequencies, such as 5.2 GHz, including the combiner 70,the LNTA 72, the filter 74, the power amplifier 64, etc., employ the InPtechnology and the components and devices that operate at the lowerfrequency, such as 1.3 GHz, employ the CMOS technology.

The modulators 52 will predominately be implemented in InP doubleheterojunction bipolar transistor (DHBT) technology where it isnecessary to meet the challenging dynamic range requirements across asignal bandwidth of 20 MHz at frequencies up to 2.6 GHz. Backendprocessing of the receive data in the DSP 54 will be implemented in40/45 nm CMOS. A combination of InP DHBT, 40/45 nm CMOS, and silicon oninsulator (SOI) 180 nm R-CMOS will be used in the transmitter module 48.The SOI CMOS supports broader frequency tuning and the InP DHBT offershigher gain for improved linearity. The DWA circuit 184 and the transmitDAC 188 will be implemented in 45 nm CMOS for highest power efficiency.

In one implementation, CMOS technology is employed to realize the datamodulation and dynamic element matching algorithms to achieve a lowpower realization. InP HBT technology is implemented in the DACs for ahigh speed operation and InP HBT or GaN HEMT technologies are employedto realize the power amplifier 64 for a large output power with highefficiency. In addition, the interleaving sigma-delta modulators can usethe micro-bump technology discussed below.

A number of technologies allow the integration of the InP fabricationtechniques and the CMOS fabrication techniques. A first enablingtechnology is an InP HBT technology having a 60 GHz FMAX that issufficient only for channelized amplifier design. A second enablingtechnology is referred to as micro-bump integration technology thatleverages CMOS and III-V material integration for mega-pixel imagingapplications. InP HBT and RF-SOI CMOS fabrication can proceed inparallel without modification and represents the best cost value forhigh voltage CMOS technology and low parasitic SOI substrates.

FIG. 13 is a profile view of a semiconductor device 280 showing anintegration between InP and CMOS technologies through a known micro-bumpintegration technique. The device 280 includes an InP substrate 282 onwhich is deposited InP device layers 284 and a CMOS substrate 286. Ametal contact layer 288 is deposited on the device layers 284 and ametal contact layer 290 is deposited on the CMOS substrate 286. Thesubstrates 282 and 286 are integrated together through a micro-bump 10μm I/O pad layers 292 and 294 having a 20 μm pad pitch.

As will be well understood by those skilled in the art, the several andvarious steps and processes discussed herein to describe the inventionmay be referring to operations performed by a computer, a processor orother electronic calculating device that manipulate and/or transformdata using electrical phenomenon. Those computers and electronic devicesmay employ various volatile and/or non-volatile memories includingnon-transitory computer-readable medium with an executable programstored thereon including various code or executable instructions able tobe performed by the computer or processor, where the memory and/orcomputer-readable medium may include all forms and types of memory andother computer-readable media.

The foregoing discussion disclosed and describes merely exemplaryembodiments of the present invention. One skilled in the art willreadily recognize from such discussion and from the accompanyingdrawings and claims that various changes, modifications and variationscan be made therein without departing from the spirit and scope of theinvention as defined in the following claims.

What is claimed is:
 1. A transceiver front-end circuit for a cellularradio, said transceiver circuit comprising: an antenna structureoperable to transmit signals and receive signals; a multiplexer coupledto the antenna structure and including multiple signal paths, eachsignal path including a bandpass filter that passes a differentfrequency band than the other bandpass filters and a circulator thatprovides signal isolation between the transmit signals and the receivesignals; a receiver module including a separate signal channel for eachof the signal paths in the multiplexer, each signal channel in thereceiver module including a receiver delta-sigma modulator that convertsanalog receive signals to a representative digital signal, whereinportions of receiver module are fabricated with indium phosphide (InP)technologies and portions of the receiver module are fabricated withCMOS technologies; and a transmitter module including a transmitterdelta-sigma modulator for converting digital data bits to the transmitsignals, said transmitter module including a power amplifier and aswitch for directing the transmit signals to one of the signal paths inthe multiplexer, wherein portions of transmitter module are fabricatedwith InP technologies and portions of the transmitter module arefabricated with CMOS technologies.
 2. The transceiver circuit accordingto claim 1 wherein components and devices that operate at higherfrequencies employ the InP technologies and components and devices thatoperate at lower frequencies and employ the CMOS technologies.
 3. Thetransceiver circuit according to claim 1 wherein at least some of theInP and CMOS technologies in the receiver module are integrated using amicro-bump integration fabrication process and at least some of the InPand CMOS technologies in the transmitter module are integrated using amicro-bump integration fabrication process.
 4. The transceiver circuitaccording to claim 1 wherein the InP technologies include InP doubleheterojunction bipolar transistor (DHBT) technologies.
 5. Thetransceiver circuit according to claim 1 wherein each receiverdelta-sigma modulator includes a combiner, a low noise amplifier (LNA),an LC filter and a quantizer circuit, said combiner receiving thereceive signals from the circulator and a feedback signal from thequantizer circuit and providing an error signal to the LNA to provide anamplified error signal, said amplifier error signal being provided tothe LC filter to provide a filtered error signal, and the filtered errorsignal being provided to the quantizer circuit, wherein the combiner,the LNA and portions of the LC filter are fabricated with InPtechnologies and portions of the LC filter are fabricated with CMOStechnologies.
 6. The transceiver circuit according to claim 5 wherein aninductor circuit in the LC filter is fabricated with InP technologiesand a capacitor circuit in the LC circuit is fabricated with CMOStechnologies.
 7. The transceiver circuit according to claim 5 whereinthe LC filter is a sixth-order filter.
 8. The transceiver circuitaccording to claim 7 wherein the LC filter includes a plurality of LCresonator circuits, a plurality of transconductance amplifiers and aplurality of integrator circuits, where a combination of one resonatorcircuit, transconductance amplifier and integrator circuit represents atwo-order stage of the LC filter.
 9. The transceiver circuit accordingto claim 8 wherein each LC circuit includes an inductor and a capacitorarray where the capacitor array includes a plurality of capacitorscontrolled by switches that provide coarse and fine tuning.
 10. Thetransceiver circuit according to claim 5 wherein the LC filter includesa low-speed digital-to-analog converter (DAC) array that receivescoefficient control bits to control the integrator circuits.
 11. Thetransceiver circuit according to claim 1 wherein the transmitter modulefurther includes a data weighted averaging (DWA) circuit that receivesthe transmit signals from the transmitter delta-sigma modulator and adigital-to-analog converter (DAC) that receives the transmit signalsfrom the DWA circuit, wherein the DAC is fabricated in InP HBTtechnologies.
 12. The transceiver circuit according to claim 11 whereinthe DWA circuit modulates the digital thermal codes to shape out voltageand timing mismatch through DAC weighting elements, and wherein the DACis a 4-bit DAC.
 13. The transceiver circuit according to claim 11wherein the transmitter delta-sigma modulator includes a dynamic elementmatching (DEM) circuit that employs an interleaving DEM algorithm. 14.The transceiver circuit according to claim 13 wherein a separate DEMcircuit is provided for each bit of the DAC.
 15. The transmitter circuitaccording to claim 1 wherein the power amplifier is fabricated in indiumphosphide (InP) heterojunction bipolar transistor (HBT) and galliumnitride (GaN) high electron mobility transistor (HEMT) technologies. 16.A transceiver front-end circuit for a cellular radio, said transceivercircuit comprising: an antenna structure operable to transmit signalsand receive signals; a multiplexer coupled to the antenna structure andincluding multiple signal paths, each signal path including a bandpassfilter that passes a different frequency band than the other bandpassfilters and a circulator that provides signal isolation between thetransmit signals and the receive signals; a receiver module including aseparate signal channel for each of the signal paths in the multiplexer,each signal channel in the receiver module including a receiverdelta-sigma modulator that converts analog receive signals to arepresentative digital signal, wherein portions of the receiver moduleare fabricated with indium phosphide (InP) technologies and portions ofthe receiver module are fabricated with CMOS technologies, and whereineach receiver delta-sigma modulator includes a combiner, a low noiseamplifier (LNA), an LC filter and a quantizer circuit, said combinerreceiving the receive signals and a feedback signal from the quantizercircuit and providing an error signal to the LNA to provide an amplifiederror signal, said amplifier error signal being provided to the LCfilter to provide a filtered error signal, and the filtered error signalbeing provided to the quantizer circuit, wherein the combiner, the LNAand portions of the LC filter are fabricated with InP technologies andportions of the LC filter are fabricated with CMOS technologies; and atransmitter module including a transmitter delta-sigma modulator forconverting digital data bits to the transmit signals, said transmittermodule including a power amplifier and a switch for directing thetransmit signals to one of the signal paths in the multiplexer, whereinportions of transmitter module are fabricated with InP technologies andportions of the transmitter module are fabricated with CMOStechnologies.
 17. The transceiver circuit according to claim 16 whereincomponents and devices that operate at higher frequencies employ the InPtechnologies and components and devices that operate at lowerfrequencies and employ the CMOS technologies.
 18. The transceivercircuit according to claim 16 wherein at least some of the InP and CMOStechnologies in the receiver module are integrated using a micro-bumpintegration fabrication process and at least some of the InP and CMOStechnologies in the transmitter module are integrated using a micro-bumpintegration fabrication process.
 19. A transceiver front-end circuit fora cellular radio, said transceiver circuit comprising: an antennastructure operable to transmit signals and receive signals; amultiplexer coupled to the antenna structure and including multiplesignal paths, each signal path including a bandpass filter that passes adifferent frequency band than the other bandpass filters and acirculator that provides signal isolation between the transmit signalsand the receive signals; a receiver module including a separate signalchannel for each of the signal paths in the multiplexer, each signalchannel in the receiver module including a receiver delta-sigmamodulator that converts analog receive signals to a representativedigital signal, wherein portions of the receiver module are fabricatedwith indium phosphide (InP) technologies and portions of the receivermodule are fabricated with CMOS technologies, and wherein each receiverdelta-sigma modulator includes a combiner, a low noise amplifier (LNA),a summation node, an LC filter and a quantizer circuit, said summationnode receiving amplified receive signals from the LNA and a feedbacksignal from the quantizer circuit and providing an error signal to theLC filter to provide a filtered error signal, where the filtered errorsignal is provided to the quantizer circuit, wherein the combiner, theLNA and portions of the LC filter are fabricated with InP technologiesand portions of the LC filter are fabricated with CMOS technologies; anda transmitter module including a transmitter delta-sigma modulator forconverting digital data bits to the transmit signals, said transmittermodule including a power amplifier and a switch for directing thetransmit signals to one of the signal paths in the multiplexer, whereinportions of transmitter module are fabricated with InP technologies andportions of the transmitter module are fabricated with CMOStechnologies.
 20. The transmitter circuit according to claim 19 whereinthe power amplifier is fabricated in indium phosphide (InP)heterojunction bipolar transistor (HBT) and gallium nitride (GaN) highelectron mobility transistor (HEMT) technologies.